DTV signal with GCR components in plural-data-segment frame headers and receiver apparatus for such signal

ABSTRACT

A DTV signal has a plural-data-segment frame header including complementary first and second ghost-cancellation reference signals differentially delayed by the duration of two NTSC horizontal scan lines. A receiver for such DTV signal has an adaptive equalizer for baseband symbol code with kernel weights calculated by a computer operative on the equalizer response after comb filtering. The comb filter combines the equalizer response as differentially delayed by the duration of two NTSC horizontal scan lines to cancel artifacts of co-channel NTSC interference in the comb filtered equalizer response supplied to the computer. This reduces the undesirable influence of such artifacts on equalization to reduce intersymbol interference.

This application is a 371 of PCT/US99/10290 filed May 11, 1999, whichclaims benefit of Ser. No. 60/085,064 filed May 12, 1998, which claimsbenefit of Ser. No. 60/089,882 filed Jun. 19, 1998, which claims benefitof Ser. No. 60/103,470 filed Oct. 8, 1998, which claims benefit of Ser.No. 60/120,638 filed Feb. 18, 1999.

The invention relates to ghost-cancellation circuitry in televisionreceivers and to reference signals included in transmitted televisionsignals for facilitating such ghost-cancellation.

BACKGROUND OF THE INVENTION

Distortion in the baseband signals recovered by a receiver caused bymulti-path reception is a problem in digital television (DTV)transmissions as well as in NTSC analog television transmissions,although the distortion is not seen as ghost images by the viewer of theimage televised by DTV. Rather the distortion causes errors in thedata-slicing procedures used to convert symbol coding to binary codegroups. If these errors are too frequent in nature, the error correctioncapabilities of the DTV receiver are overwhelmed, and there iscatastrophic failure in the television image. If such catastrophicfailure occurs infrequently, it can be masked to some extent by freezingthe good TV images most recently transmitted, such masking being lesssatisfactory if the TV images contain considerable motion content. DTVreceivers use adaptive equalizers to suppress the distortion caused bymultipath reception, which equalizers are similar to those previouslyused in some NTSC television receivers. The adaptive equalizers aredigital filters with kernel weights that can be adjusted by suitableelectronics to reduce multi-path signals known as “pre-ghosts” that arereceived before the principal signal is received and to reducemulti-path signals known as “post-ghosts” that are received after theprincipal signal is received.

Several forms of adaptive equalizer are known. The adaptive equalizercan be a finite-impulse-response (FIR) digital filter formed from aseveral-bit-wide digital shift register a few hundred stages in lengthand a respective 4-quadrant digital multiplier for each stage to weightthe contents of that stage by respective kernel weight for inclusion ina weighted summation. However, since many of the kernel weights are ofnegligible value, such a straightforward approach is wasteful of digitalhardware. Adaptive equalizers currently preferred by many personsskilled in the art incorporate cascades of digital filters withspecialized functions, such as the cancellation of pre-ghosts occurringa substantial number of microseconds before the principal signal, thecancellation of post-ghosts occurring a substantial number ofmicroseconds after the principal signal, and the cancellation ofso-called “micro-ghosts” that occur close in time to the principalsignal but affect the amplitude and phase characteristics of theprincipal signal in undesired degree. The two types of digital filterearlier referred to are sometimes referred to as “ghost cancellationfilters” in contradistinction to the last of these types of digitalfilter with specialized function being sometimes referred to as an“equalizer”, but in this specification the term “equalizer” is used in ageneric sense to include all these species of digital filter.

A standard for digital high-definition television (HDTV) signalspublished 16 Sep. 1995 by the Advanced Television Systems Committee(ATSC) is currently accepted as the de facto standard for terrestrialbroadcasting of digital television (DTV) signals in the United States ofAmerica. The data is transmitted in a succession of consecutive-in-timedata fields each containing 313 consecutive-in-time data segments ordata lines. The data is randomized and interleaved with a52-data-segment (inter-segment) convolutional byte interleaver duringits arrangement into the data fields. Each segment of data is precededby a data segment synchronization code group of four symbols havingsuccessive values of +S, −S, −S and +S. The value +S is one level belowthe maximum positive data excursion, and the value −S is one level abovethe maximum negative data excursion. The segments of data are each of77.3 microsecond duration, and there are 832 symbols per data segmentfor a symbol rate of about 10.76 MHz. The initial line of each datafield is a data field synchronization (DFS) code group that codes atraining signal for channel-equalization and multipath suppressionprocedures. The remaining lines of each data field contain data thathave been Reed-Solomon forward error-correction coded. In over-the-airbroadcasting the error-correction coded data are then trellis codedusing twelve interleaved trellis codes, each a 2/3 rate puncturedtrellis code with one uncoded bit. Trellis coding results are parsedinto three-bit groups for over-the-air transmission in eight-levelone-dimensional-constellation symbol coding, which transmission is madewithout symbol pre-coding separate from the trellis coding procedure.Trellis coding is not used in cablecasting proposed in the ATSCstandard. The error-correction coded data are parsed into four-bitgroups for transmission as sixteen-level one-dimensional-constellationsymbol coding, which transmissions are made without precoding.

The carrier frequency of a VSB DTV signal is 310 kHz above the lowerlimit frequency of the TV channel. The VSB signals have their naturalcarrier wave, which would vary in amplitude depending on the percentageof modulation, suppressed. The natural carrier wave is replaced by apilot carrier wave of fixed amplitude, which amplitude corresponds to aprescribed percentage of modulation. This pilot carrier wave of fixedamplitude is generated by introducing a direct component shift into themodulating voltage applied to the balanced modulator generating theamplitude-modulation sidebands that are supplied to the filter supplyingthe VSB signal as its response. If the eight levels of 4-bit symbolcoding have normalized values of −7, −5, −3, −1, +1, +3, +5 and +7 inthe carrier modulating signal, the pilot carrier has a normalized valueof 1.25. The normalized value of +S is +5, and the normalized value of−S is −5.

In the ATSC standard published 16 Sep. 1995 the data fieldsynchronization signals in initial data segments of consecutive datafields were designed for use as ghost-cancellation reference (GCR)signals to train adaptive equalization circuitry in the DTV signalreceiver. The training signal or GCR signal in the initial data segmentof each data field is a 511-sample pseudo-random noise sequence referredto as “PN511 signal” followed by three 63-sample pseudo-random noisesequences referred to as “PN63 signals”. The middle ones of the PN63signals in the field synchronization codes are transmitted in accordancewith a first logic convention in the first of 313 data segments in eachodd-numbered data field and in accordance with a second logic conventionin the first of 313 data segments in each even-numbered data-field, thefirst and second logic conventions being one's complementary respectiveto each other. The other two PN 63 signals and the PN511 signal aretransmitted in accordance with the first logic convention in each andevery data field.

The middle PN63 sequence of the ATSC field synchronization code, asseparated by differentially combining corresponding samples ofsuccessive field synchronization code sequences, can used as a basis fordetecting ghosts. Pre-ghosts of up to −47.848 microseconds (578 symbolperiods) before the separated middle PN63 sequence can be detected in adiscrete Fourier transform (DFT) procedure without having todiscriminate against data in the last data segment of the preceding datafield. However, the post-ghosts of such data can extend up to fortymicroseconds into the first data segments and add to the backgroundclutter that has to be discriminated against when detecting pre-ghostsof the separated middle PN63 sequence. Post-ghosts of up to 18.117microseconds (195 symbol periods) after the separated middle PN63sequence can be detected in a DFT procedure without having todiscriminate against data in the precode and in the data segment of thesucceeding data field. Longer-delayed post-ghosts have to be detectedwhile discriminating against background clutter that includes data. Theauto-correlation properties of the PN63 sequence are not so great thatdetection of longer-delayed post-ghosts is sufficiently sensitive, itappears in practice. The middle PN63 sequence of the ATSC fieldsynchronization code provides more pre-ghost canceling capability thanrequired in practice, but insufficient post-ghost canceling capability.While post-ghosts delayed up to forty microseconds after principalsignal occur in actual practice, pre-ghosts advanced more than sixmicroseconds before principal signal do not occur except in a poorlyshielded TV receiver where a signal may be received by direct radiationas much as thirty microseconds before the same signal received viacable. Pre-ghosts preceding the principal signal by more than fourmicroseconds are rare, according to page 3 of the T3S5 Report GhostCanceling Reference Signals published 20 Mar. 1992 by the ATSC.

Modifying the ATSC field synchronization code, so as to place the threePN63 sequences immediately after the 4-symbol segment synchronizationcode and an immediately pursuant 24-symbol VSB-mode code, to be followedby the PN511 sequence and the 104-symbol gap referred to as “reserve”,would improve the ghost-separation capabilities of the separated middlePN63 sequence. Post-ghosts up to 63.364 microseconds (682 symbolsduration) and pre-ghosts up to −8.455 microseconds (91 symbols duration)could be detected without data making substantial contribution tobackground clutter.

If one seeks to exploit the auto-correlation properties of the PN511sequence in the ATSC DTV signal for selection of ghosts in a DFTprocedure, the selection filter has to discriminate PN511 sequence andits ghosts from background clutter that includes data and the initialand final PN63 sequences. This background clutter has substantialenergy, so weaker ghosts of the PN511 sequence are difficult to detect.The higher energy response of the PN511 auto-correlation filter used forghost detection cannot be fully exploited because data and the initialand final PN63 sequences increase so much the energy of the backgroundclutter that the filter is to discriminate against.

The training signal or GCR signal is used in many adaptive equalizersjust for initializing the kernel weights, since initialization can becarried out more rapidly than is the case if adjustments of the kernelweights are made based on decisions as to the data content of currentlyreceived signal. Also, initialization using a training signal avoids thepossibility of adjustments of the kernel weights stalling duringleast-mean-squares error calculations when a localized optimization ofkernel weights is achieved that is not an ultimate optimization of thosekernel weights. After initialization is accomplished, adjustments of thekernel weights are better made based on decisions as to the data contentof currently received signal in a decision-directed adjustment procedurethat can more rapidly adjust to changes in multi-path receptionconditions, to permit tracking those changes sufficiently well thatcorruption of received signal is not so great as to cause frequent errorin determining its data content. The need to acquire the training signalor GCR signal over the course of several data fields, in order tosuppress ghosts of the principal signal sufficiently thatdecision-directed adjustment procedures can take over, presents aproblem with ever being able initially to establish tracking of changingmulti-path reception conditions.

This problem is avoided in accordance with an aspect of the invention byacquiring the GCR signal in a plurality of consecutive data segmentswithin each data frame of the DTV signal. Acquiring GCR signal inconsecutive horizontal trace intervals of an NTSC television signal isinfeasible because: the information content of those lines is used tocontrol quite directly the image traced onto the viewscreen duringvertical trace period, there is no stable clock source (i.e., colorburst) during the earlier horizontal trace intervals within verticalretrace period, and as a practical matter all but one of the laterhorizontal trace intervals within vertical retrace period is bespokenfor other uses. These reasons are inapplicable to acquiring GCR signalin consecutive data segments of the DTV signal, or can be made so. Thereis no conformal mapping of the received data and the image traced on theviewscreen, so how GCR signal is inserted into the data stream islargely a matter of choice as long as data buffering requirements in theinformation pipeline do not become excessive. Measures can be providedfor stabilizing the timing of sample clocks that are harmonicallyrelated to regenerated symbol clock during the consecutive data segmentsof the DTV signal in which the GCR signal is acquired. Since DTV is anew technology, there are no previous commercial considerationsgoverning the use of particular data segments in the signal.

Another problem with the ghost suppression techniques previouslyattempted in DTV is that the effects of co-channel NTSC interferencehave not been given sufficient attention. The GCR signal received by aDTV signal receiver can be contaminated by artifacts of co-channel NTSCinterference, particularly by standing frequencies associated with NTSCvideo carrier, chroma subcarrier and audio carrier. The adaptiveequalizer will attempt to diminish response at those standingfrequencies, which undesirably affects symbol decoding procedures. Theartifacts of co-channel NTSC audio signal interference can be avoided byselective filtering in the intermediate-frequency amplifier of the DTVsignal receiver, as described by A. L. R. Limberg in U.S. patentapplication Ser. No. 08/826,790 filed 24 Mar. 1997, entitled “DTVRECEIVER WITH FILTER IN I-F CIRCUITRY TO SUPPRESS FM SOUND CARRIER OFNTSC CO-CHANNEL INTERFERING SIGNAL”, and incorporated herein byreference.

In accordance with an aspect of the invention received DTV basebandsignal and that signal as delayed 1368 symbol epochs (the duration oftwo NTSC horizontal scan lines) are subtractively combined in a combfilter supplying a response from which training signal is to beextracted. The artifacts of co-channel NTSC video carrier and chromasubcarrier are eliminated in this comb filter response. The datasegments of the DTV signal from which the GCR signal and its ghosts areto be acquired are designed each to include the GCR signal and itscomplement delayed 1368 symbol epochs (the duration of two NTSChorizontal scan lines) from the original GCR signal. Accordingly, thisGCR signal is reproduced with doubled energy in the response of the combfilter used to suppress the artifacts of co-channel NTSC video carrierand chroma subcarrier.

SUMMARY OF THE INVENTION

An aspect of the invention concerns an electromagnetic wave signalcomprising vestigial sideband modulation of a suppressed carrier inaccordance with a baseband signal having a uniform symbol ratesubstantially 684 times the horizontal scan line rate of an NTSCtelevision signal that is apt to accompany the electromagnetic wavesignal as a co-channel interfering signal, the baseband signal beingcomposed of consecutive data segments each consisting of a prescribedintegral number of symbol epochs, and the consecutive data segmentsbeing divided into contiguous data frames each consisting of aprescribed integral number M of contiguous ones of those data segments.In an electromagnetic wave signal embodying this aspect of theinvention, each data frame begins with a plurality N in number of datasegments used as a frame header and concludes with a plurality (M-N) innumber of data segments composed of consecutive multi-level symbols usedfor transmitting data. The frame header includes a firstghost-cancellation reference signal and a second ghost-cancellationreference signal at a prescribed time interval thereafter, whichprescribed time interval is different than the duration of a datasegment. The first and second ghost-cancellation reference signalexhibit respective variations which are of opposite sense to eachother—i.e., which are complementary to each other. Preferably, the firstghost-cancellation reference signal comprises a plurality of PNsequences that are orthogonal to each other and contain an equal numberof symbols. Preferably, the first ghost-cancellation reference signalbegins substantially 1368 symbol epochs before the secondghost-cancellation reference signal. Other aspects of the inventionconcern baseband symbol coding corresponding to the electromagnetic wavesignals embodying the aspects of invention previously described in thisparagraph.

Still other aspects of the invention concern data signal receivers forthe electromagnetic wave signals embodying the aspects of inventiondescribed in the foregoing paragraph. Such a data signal receiverincludes circuitry for selecting one of these electromagnetic wavesignals, converting the frequencies of the electromagnetic wave signalafter its selection, and amplifying the electromagnetic wave signalafter its selection and conversion in frequency. Such a data signalreceiver includes circuitry for synchrodyning the electromagnetic wavesignal to baseband after its selection, conversion in frequency andamplification and supplying digitized samples of a baseband signalresulting from synchrodyning the electromagnetic wave signal tobaseband. These samples are supplied as input signal to an adaptiveequalizer for supplying an equalizer response to those received samplesas weighted by kernel weights that are electrically adjustable. Such adata signal receiver includes circuitry for regenerating transmitteddata from the equalizer response. A comb filter is included in the datasignal receiver for differentially delaying the equalizer response, sothe first ghost-cancellation reference signal in the more delayedequalizer response occurs simultaneously with the secondghost-cancellation reference signal in the less delayed equalizerresponse, and for subtractively combining the more delayed equalizerresponse and the less delayed equalizer response to generate a combfilter response. A computer is arranged to respond to selected portionsof the comb filter response including the result of subtractivelycombining the first and second ghost-cancellation reference signals, forperforming initial electrical adjustments of the kernel weights of theadaptive equalizer whenever the data signal receiver is initiallyoperated after a time of in operation or whenever the electromagneticwave signal is initially selected.

Preferably, during continued operation of the data signal receiver, thecomputer electrically adjusts the kernel weights of the adaptiveequalizer responsive to the comb filter response on a decision-directedbasis. Alternatively, the computer may continue to update itsadjustments of the kernel weights of the adaptive equalizer responsiveto the comb filter response by continuing to use as a training signalthose portions of the comb filter response including the result ofsubtractively combining the first and second ghost-cancellationreference signals.

BRIEF DESCRIPTION OF THE DRAWING

FIG. 1 is a block schematic diagram of a vestigial-sideband digitaltelevision signal receiver embodying the invention in one of itsaspects.

FIG. 2 is a block schematic diagram of modifications of the FIG. 1vestigial-sideband digital television signal receiver that result inanother vestigial-sideband digital television signal receiver, whichembodies the invention in one of its aspects.

FIGS. 3A, 3B, 3C and 3D are timing diagrams showing the nature of thesymbol coding in a data frame header consisting of the first four datasegments L₁, L₂, L₃, and L₄ of each data frame of a vestigial-sidebanddigital television signal embodying aspects of the invention.

FIGS. 4A, 4B, 4C and 4D are timing diagrams showing the nature of thesymbol coding in a data frame header consisting of the first four datasegments L₁, L₂, L₃, and L₄ of each data frame of another VSB DTV signalembodying aspects of the invention.

FIGS. 5A, 5B, 5C and 5D are timing diagrams showing the nature of thesymbol coding in a data frame header consisting of the first four datasegments L₁, L₂, L₃, and L₄ of each data frame of another VSB DTV signalembodying aspects of the invention.

FIGS. 6A, 6B, 6C and 6D are timing diagrams showing the nature of thesymbol coding in a data frame header consisting of the first four datasegments L₁, L₂, L₃, and L₄ of each data frame of another VSB DTV signalembodying aspects of the invention.

FIGS. 7A, 7B, 7C and 7D are timing diagrams showing the nature of thesymbol coding in a data frame header consisting of the first four datasegments L₁, L₂, L₃, and L₄ of each data frame of another VSB DTV signalembodying aspects of the invention.

FIGS. 8A, 8B, 8C and 8D are timing diagrams showing the nature of thesymbol coding in a data frame header consisting of the first four datasegments L₁, L₂, L₃, and L₄ of each data frame of another VSB DTV signalembodying aspects of the invention.

FIGS. 9A, 9B, 9C and 9D are timing diagrams showing the nature of thesymbol coding in a data frame header consisting of the first four datasegments L₁, L₂, L₃, and L₄ of each data frame of another VSB DTV signalembodying aspects of the invention.

FIG. 10 is a diagram of a 1252-data-segment data frame of less than 100millisecond duration, which data frame includes a frame headerconsisting of the four data segments L₁, L₂, L₃, and L₄, each being 832symbol epochs in duration.

FIG. 11 is a diagram of a data frame of slightly more than 100millisecond duration, which data frame includes a frame headerconsisting of the four data segments L₁, L₂, L₃, and L₄, each being 832symbol epochs in duration.

FIG. 12 is a diagram of a data frame of slightly more than 100millisecond duration, which data frame includes a frame headerconsisting of the four data segments L₁, L₂, L₃, and L₄, each being 828symbol epochs in duration.

DETAILED DESCRIPTION

In FIG. 1 antenna 1 represents all sources of radio-frequency (RF)television signals to the receiver shown therein. The FIG. 1 VSB DTVsignal receiver is used for recovering error-corrected data in packetform, which packets are suitable for recording by a digital videocassette recorder or for decoding and presentation in a television set.The FIG. 1 receiver includes a radio-frequency (RF) amplifier 2 foramplifying a selected RF signal for application to first detectorcircuitry 3 for conversion to a first intermediate-frequency (I-F)signal. The first IF signal is in an ultra-high-frequency (UHF)intermediate-frequency band located above 890 MHz, the upper limitfrequency of channel 83, the highest frequency ultra-high-frequency TVbroadcast channel. The RF amplifier 2 has a bandpass filter that limitsthe frequency range of radio-frequency input signals to the 50 to 890MHz band, rejecting image frequencies above the first IF band. The RFamplifier 2 also includes a broadband tracking filter that rejectstelevision signals other than that selected for viewing, especiallythose signals of higher power such as analog TV signals innext-to-adjacent channels. The first detector 3 includes a first localoscillator for generating first local oscillations tunable over a rangefrom 970 to 1730 MHz to place the first IF signal in a 6-MHz-wide bandcentered at about 920 MHz with the carrier nominally at 922.69 MHz.These first local oscillator frequencies are such that any leakage fromthe tuner input will not interfere with UHF TV channels as may bereceived by another television signal receiver nearby. At the same timesecond harmonics of UHF TV channels 14 through 69 fall above the firstIF band. A first mixer included in the first detector 3 is a highlylinear doubly-balanced type to avoid even harmonic generation, and thereis a broadband tuned coupling for rejecting image frequencies in thefirst IF signal supplied from the first detector 3 to a firstintermediate-frequency amplifier 4.

The first IF amplifier 4 is sometimes referred to as the “UHFintermediate-frequency amplifier” or “UHF IF amplifier” since itprovides amplification of signals in an ultra-high-frequency first IFband above the UHF television broadcasting channels. The first IF signalexhibits a frequency spectrum reversal relative to the RF signalsupplied to the first mixer in the first detector 3. The first IFamplifier 4 provides constant, linear gain to overcome the 10–12 dBinsertion loss in a first surface-acoustic-wave filter included therein.This first SAW filter can be constructed on a gallium arsenidesubstrate. The constant gain makes it simpler for to drive the SAWfilter at all times from the source impedance prescribed for avoidingmultiple reflections that interfere with obtaining good group delay.Since its gain is not automatically adjusted, the first IF amplifier 4is designed to have as much dynamic range as possible to avoidcross-modulation of co-channel interfering analog TV signals with DTVsignals.

The first IF amplifier 4 response is supplied as input signal to asecond detector 5, there to be mixed with second local oscillations togenerate DTV signals translated to a very-high-frequency second IF bandbelow the VHF television broadcasting channels. There is a broadbandtuned coupling for rejecting image frequencies in output signal from thesecond detector 5, which can be provided by a second SAW filter. Thissecond SAW filter is typically constructed on a lithium niobatesubstrate. Supposing that the response of the first SAW filter in firstIF amplifier 4 is relied upon to define overall IF bandwidth, the secondSAW filter has an amplitude response that is substantially flat overthat bandwidth and exhibits a phase response that is substantiallylinear. If prior art practice is followed, second local oscillationshaving a frequency of 876 MHz are used to supply a second mixer in thesecond detector 5 and the downconversion result resides in a secondintermediate-frequency band located in the 41–47 MHz frequency rangeconventionally used as the intermediate frequency band in analog TVsignal receivers of single-conversion type. The first IF band may bedisplaced somewhat from 917–923 MHz and the second local oscillationsmay be at a frequency either below or above the first IF band infrequency. If the second local oscillations are below the first IF bandin frequency, like the first IF signal the second IF signal suppliedfrom the second detector 5 exhibits a frequency spectrum reversalrelative to the RF signal supplied to the first detector 3. If thesecond local oscillations are above the first IF band in frequency, thesecond IF signal exhibits a frequency spectrum that is not reversedrelative to the RF signal supplied to the first detector 3.

The second detector 5 response is supplied as input signal to a secondintermediate-frequency amplifier 6. The second IF amplifier 6 issometimes referred to as the “VHF intermediate-frequency amplifier” or“VHF IF amplifier” since it provides amplification of signals in avery-high-frequency second IF band below the VHF television broadcastingchannels. The second IF amplifier 6 has a plurality of cascadedamplifier stages, which are controlled in gain as part of a delayedautomatic-gain-control (AGC) system. An amplifier controlled in gainusing AGC of reverse type generally has poorer noise figure than anamplifier controlled in gain using AGC of forward type in whichamplifier stages are operated partially in a saturated condition,however. The noise figures of the stages in the second IF amplifier 6are of less concern, owing to their later position in the amplifierchain, so these amplifier stages are preferably controlled in gain usingAGC of reverse type better to keep non-linearity within reasonablelimit. Controlling the gain of the second IF amplifier 6 using AGC ofreverse type also makes it easier to prevent problems of phase shift asa function of change in modulation levels. In an analog TV receiverearlier IF amplifiers are preferably controlled in gain using AGC offorward type so that the video signal is free of noise that appears as“snow”, particularly “color snow”, on the television viewscreen. As longas noise is smaller than the smallest modulation steps in the DTVsignal, so that the noise does not “capture” the data-slicing proceduresused in symbol decoding, the presence of noise is of little consequencein a DTV signal receiver. The quantizing effects of the data-slicingprocedures used in symbol decoding suppress the effects of noise untilit exceeds the smallest modulation steps in the DTV signal. Infrequentbursts of noise that exceed the smallest modulation steps in the DTVsignal can be corrected by the decoding of trellis coding andReed-Solomon forward-error correction coding which decoding is performedlater on in the VSB DTV signal receiver.

The second IF amplifier 6 response is supplied as input signal to athird mixer 7, there to be mixed with third local oscillations from acontrolled third local oscillator 8. The local oscillator 8 and themixer 7 function as a third detector to generate VSB DTV signaltranslated in frequency to a third IF band within a 1–10 MHz range. Thislower intermediate frequency facilitates digitization of the VSB DTVsignal by an analog-to-digital converter 9. Digitization is done at amultiple at least two of the symbol rate of the VSB DTV signal. Thedigitized VSB DTV signal is synchrodyned to baseband in circuitry 10that comprises: filters for converting the digitized samples to complexform, read-only memory (ROM) storing sine and cosine lookup tables for acomplex digital carrier, and a digital multiplier for complex numbersmultiplying the result of converting digitized samples to complex form(as multiplicand) by the complex digital carrier (as multiplier) togenerate a complex product comprising a stream of digital samples of thereal portion of the baseband symbol coding and a parallel stream ofdigital samples of the imaginary portion of the baseband symbol coding.The stream of digital samples of the real portion of the baseband symbolcoding and in some designs the stream of digital samples of theimaginary portion of the baseband symbol coding are supplied to anadaptive channel-equalization filter or equalizer 11.

A digital-to-analog converter 12 converts the digital samples of theimaginary portion of the baseband symbol coding to an analog errorsignal lowpass filtered by an automatic-frequency-and-phase controlfilter 13 to develop an automatic-frequency-and-phase control (AFPC)signal for the controlled third local oscillator 8, to complete anautomatic-frequency-and-phase control (AFPC) feedback loop. The AFPCloop minimizes the low-frequency energy in the digital samples of theimaginary portion of the baseband symbol coding, which should be zero inbaseband symbol coding of a VSB DTV signal, and maximizes thelow-frequency energy in the digital samples of the real portion of thebaseband symbol coding.

The digital samples of the real portion of the equalized baseband symbolcoding from the equalizer 11 are supplied to a symbol decoder 14. Thesymbol decoder 14 customarily includes data slicer circuitry and aViterbi decoder and often includes comb filtering to reject co-channelNTSC interference. The symbol decoding results are supplied to trellisdecoder circuitry 15, which includes a respective trellis decoder andpost coder for each of the independent codestreams in the VSB DTVsignal, there being twelve such independent codestreams in the VSB DTVsignal specified in the ATSC standard published 16 Sep. 1995. Thetrellis decoder circuitry 15 supplies its response to a de-interleaver16 to be reformatted into successive 8-bit bytes and to undo theseveral-data-segment convolutional byte interleaving done at thetransmitter. Burst errors in the trellis coding are dispersed in thede-interleaved forward error-corrected code bytes supplied toReed-Solomon decoder circuitry 17 for correction of errors. TheReed-Solomon decoder circuitry 17 supplies the error-corrected data tode-randomizer circuitry 18 to undo the data randomization performed atthe transmitter. The recovered data are then supplied from thede-randomizer circuitry 18 to a packet sorter 19. The packet sorter 19selects packets of video information and packets of audio informationfor use in the remainder of the DTV signal receiver, which may comprisedisplay and speaker components of a television set, or which mayalternatively comprise the recording electronics of a digital recorder.

Sample clock generation circuitry 20 generates sampling clock signalsthat are harmonically related to the symbol rate of the received VSB DTVsignal, preferably using a passband spectral-line timing recoverytechnique for sample clock phase synchronization. An envelope detector21 detects the envelope of amplified VHF IF signal from the I-Famplifier 6. A narrow bandpass filter 22 selects the components ofbaseband signal occurring in the envelope detector 21 response atone-half baud rate (i.e. one-half symbol frequency). Circuitry 23 forsynchronizing the sample clock with the symbol rate customarilycomprises a frequency multiplier, a controlled oscillator, a clockedfrequency divider and a phase detector, none of which elements areexplicitly shown in FIG. 1. The frequency multiplier doubles orquadruples the frequency of the half-symbol-frequency response of thefilter 22 to generate a frequency multiplier response. The clockedfrequency divider divides the frequency of the oscillations from thecontrolled oscillator to generate a frequency divider nominally of thesame frequency as the frequency multiplier response. An automaticfrequency and phase control (AFPC) signal for the controlled oscillatoris supplied from the phase detector which compares the phasing of thefrequency divider response with the phasing of the frequency multiplierresponse. The controlled oscillator is preferably crystal controlled, sothe AFPC needs primarily to make adjustments to phase. The controlledoscillator generates oscillations, the average-axis-crossings of whichtime the edges of sample clock signals.

The sample clock generation circuitry 20 supplies clocking signals tothe ADC 9 and to the digital synchrodyne circuitry 10. These clockingsignals are supplied at a sample rate that is a multiple at least two ofsymbol rate. The ADC 9 uses these clocking signals for controlling thedigital sampling of the low final intermediate-frequency VSB DTV signal.The digital synchrodyne circuitry 10, the components of which are notexplicitly shown in FIG. 1, includes an address counter for countingthese clocking signals to generate addressing for a read-only memorythat stores look-up tables for the sine and cosine of the complexdigital carrier used for synchrodyning. The sample clock generationcircuitry 20 supplies clocking signals at symbol rate to the dataslicing circuitry in the symbol decoder 14. The sample clock generationcircuitry 20 supplies clocking signals at symbol rate to the adaptiveequalizer 11 for timing its output samples to the symbol decoder 14. Ifthe adaptive equalizer 11 is a fractional equalizer rather than asynchronous equalizer, the sample clock generation circuitry 20 suppliesfurther clocking signals to the equalizer 11 at a rate higher thansymbol rate but related in whole-number-ratio to the symbol rate. Thesample clock generation circuitry 20 supplies various clocking signalsto the trellis decoder circuitry 15, the de-interleaver circuitry 16,the Reed-Solomon decoder circuitry 17, and the de-randomizer circuitry18.

The sample clock generation circuitry 20 supplies clocking signals to asample counter 24 which counts the number of samples in a data frame andthen rolls over its count to provide a modular count. A frame startdetector 25 resets the count from the sample counter 24 to a prescribedvalue when a frame start sequence is detected in the stream of digitalsamples of the real portion of the equalized baseband symbol codingsupplied by the equalizer 11. The frame start detector 25 typicallyincludes a finite-impulse-response digital filter with a kernel thatprovides a match filter for the frame start sequence, and a thresholddetector for determining when the match filter response peaks.

A computer 26 determines the kernel weights for the digital filteringincluded in the adaptive equalizer 11. The FIG. 1 VSB DTV signalreceiver differs from those known in the art in that digital samples ofthe real portion of the equalized baseband symbol coding are notreceived directly into the computer 26, but instead are comb filtered bya comb filter 30 that suppresses artifacts of co-channel NTSCinterference that accompany these samples. The comb filter 30 includes ashift register 31 and a digital signed adder 32 operated as asubtractor. The shift register 31 is 1368 stages long, each stage beingapproximately 10–12 bits wide, and is operated as a clocked delay linefor differentially delaying digital samples of the real portion of theequalized baseband symbol coding by the duration of two NTSC horizontalscan lines. The adder 32 differentially combines the differentiallydelayed digital samples of the real portion of the equalized basebandsymbol coding to generate a difference signal containing GCR signal forapplication to the computer 26.

The general forms that the adaptive equalizer 11 and the computer 26 areapt to take are known to those skilled in the art of digitalcommunications receiver design. The general forms that the adaptiveequalizer 11 is likely to take are outlined in the BACKGROUND OFINVENTION, supra. The general form that the computer 26 preferably takesis determined in large measure by certain equalization systempreferences.

The preference is for a system in which the kernel weights for thedigital filtering are initialized by evaluating a prescribed trainingsignal time-division-multiplexed with the digital modulation at thetransmitter and received together with multi-path distortion at thereceiver. The prescribed training signal with multi-path distortion asreceived by the receiver is compared with an ideal prescribed trainingsignal free from multi-path distortion as stored at the receiver toevaluate the nature of the multipath distortion. The computer 26 hasrandom-access memory (RAM) included therein for storing segments of thestream of digital samples of the real portion of the equalized basebandsymbol coding supplied from the equalizer 11, which segments contain thereceived training signal or ghosts thereof. The computer 26 is receptiveof the sample count from the sample counter 24 to have informationconcerning when the segments of that stream of digital samples whichcontain the received training signal or ghosts thereof occur, so a WRITECOMMAND can be supplied to the RAM, which is addressed by a portion ofthe sample count from the sample counter 24 during the writing of theRAM.

Especially if the differential delays between the principal receivedsignal and its ghosts are not too long, discrete Fourier transform (DFT)methods can be used to initialize the kernel weights for the digitalfiltering in a very short time. The DFT of the prescribed trainingsignal with multi-path distortion as received by the receiver iscalculated and divided by the DFT of the prescribed training signal todetermine the DFT of the transmission channel, a process referred to as“characterizing the channel”. The complement of the channel DFTdescribes the DFT the adaptive equalizer should have, and the kernelweights are determined accordingly. These calculations are carried outby a microprocessor with suitable software being included in thecomputer 26. The computer 26 can include read-only memory (ROM) forstoring the DFT of the prescribed training signal; this saves having tocalculate the DFT of the prescribed training signal from the prescribedtraining signal per se as stored and read from ROM.

If the differential delay between the principal received signal and aghost thereof is substantial, several microseconds or tens ofmicroseconds, a match filter for the training signal can be included inthe computer 26 and used in conjunction with a microprocessor alsoincluded in the computer 26 for determining the differential delay andthe relative magnitude of the ghost. This can speed up the calculationof kernel weights for filters which use programmable bulk delay betweensparse groupings of taps to have non-zero weights

The preference is for a system in which the kernel weights for thedigital filtering after being initially determined thereafter continueto be adjusted by decision-directed methods. This permits changingmultipath conditions to be tracked on a continuous basis.Decision-directed methods are best implemented by including a companiondigital filter in the computer 26. Such a procedure using aleast-mean-squares (LMS) optimization procedure implemented on a blockbasis is described in detail in U.S. Pat. No. 5,648,987 entitled“RAPID-UPDATE ADAPTIVE CHANNEL-EQUALIZATION FILTERING FOR DIGITAL RADIORECEIVERS, SUCH AS HDTV RECEIVERS” issued 15 Jul. 1997 to J. Yang, C. B.Patel, T. Liu and A. L. R. Limberg. Such a procedure using an LMSoptimization procedure implemented on a continuous basis is described indetail in allowed U.S. patent application Ser. No. 08/832,674 entitled“DYNAMICALLY ADAPTIVE EQUALIZER SYSTEM AND METHOD” filed 8 Apr. 1997 byA. L. R. Limberg. C. M. Zhao, X. Y. Hu and X. H. Yu indicate in theirSeptember 1998 paper “Block Sequential Least Squares Decision FeedbackEqualization Algorithm with Application to Terrestrial HDTVTransmission” appearing in IEEE Transactions on Broadcasting, Vol. 44,No. 3, that using block-sequential LMS optimization procedures, ratherthan continuous LMS optimization procedures, permits a bit error rate of3×10⁻⁹ to be achieved with signals having 3.5 dB poorer signal-to-noiseratio.

FIG. 2 shows a modification of the FIG. 1 VSB DTV signal receiver inwhich the VHF IF amplifier 6 response is supplied to synchronousdetectors 71 and 72 that replace the third mixer 7. In FIG. 2 the thirdlocal oscillator 8 is replaced by a third local oscillator 81 supplyingthird local oscillations in 00 phasing to the mixer 71 and in 90°phasing to the mixer 72, the frequency of the local oscillations beingat the carrier frequency of the VSB DTV signal in the VHF IF amplifier 6response. The ADC 9 is replaced by two analog-to-digital converters 91and 92 which digitize the responses of the synchronous detectors 71 and72 with a sampling rate that is a multiple at least two of the symbolrate of the VSB DTV signal. The ADC 91 digitizes the baseband symbolcode response of the synchronous detector 71 to generate a stream ofdigital samples of the real portion of the baseband symbol codingsupplied to the equalizer 11. The ADC 92 digitizes the baseband symbolcode response of the synchronous detector 72 to generate a stream ofdigital samples of the imaginary portion of the baseband symbol codingsupplied to the digital-to-analog converter 13 and to the equalizer 11if the equalizer 11 is a complex equalizer.

Each successive pair of data frames consisting of four data fields, eachwith 313 data segments and an initial data segment containing data fieldsynchronization code, that would occur in a VSB DTV signal generated inaccordance with the ATSC standard published 16 Sep. 1995, is replaced inVSB DTV signals embodying certain aspects of the invention by a dataframe with 1252 data segments, the first four data segments providing aframe header containing data field synchronization code andghost-cancellation reference signal. VSB DTV signals that have dataframes with a different number of data segments, but still have thefirst four data segments of each data frame containing data fieldsynchronization code and ghost-cancellation reference signal, embodysimilar aspects of the invention and may prove preferable in practice.For example, data frames with 1294 or 1295 data segments make it simplerto temporally track groups of images than data frames with 1252 datasegments do.

The underlying concept of the invention in a principal one of itsaspects is that, when multi-path distortion is changing, dispersal ofghost-cancellation reference signal components into non-contiguous datasegments that are to be later collected together makes it difficult orimpossible to initialize the kernel weights in the adaptive equalizer 12in the VSB DTV signal receiver so as to open the eye for data slicing.Placing the ghost-cancellation reference signal components intocontiguous data segments reduces the effect of changing multi-pathdistortion on being able properly to combine the ghost-cancellationreference signal components, making it more likely that the kernelweights in the adaptive equalizer 12 can be initialized so as to openthe eye for data slicing sufficiently that decision-directed procedurescan then be instituted for further improving equalization and fortracking equalization to changing to initialize the kernel weights inthe adaptive equalizer 12 in the VSB DTV signal receiver so as to openthe eye for data slicing.

Arranging for a plurality data segments containing components of GCRsignal to be contiguous makes it possible to arrange those components sothat in a VSB DTV signal receiver a comb filter can combine thecomponents to suppress artifacts of co-channel NTSC interference inbaseband symbol code recovered by synchrodyne.

FIGS. 3A, 3B, 3C and 3D are timing diagrams showing one way to arrangethe symbol coding in a data frame header consisting of the first fourdata segments L₁, L₂, L₃, and L₄ of each data frame of avestigial-sideband digital television signal such that the comb filter30 in the VSB DTV signal receiver of FIG. 1 or in such a receivermodified per FIG. 2 will recover ghost-cancellation reference signal inwhich co-channel NTSC interference is suppressed. In a VSB DTV receiverthe four data segments L₁, L₂, L₃, and L₄ as recovered by synchrodyningthe DTV signal to baseband are superposed on a direct componentattributable to synchronous detection of the pilot carrier. This directcomponent is omitted in all the timing diagrams in the drawing, so thattheir remaining content is more readily understood. Furthermore, thisdirect component cancels in the response of the comb filter 30 and doesnot affect computations of kernel weights by the computer 29.

In FIGS. 3A, 3B, 3C and 3D the four data segments L₁, L₂, L₃, and L₄begin with data segment sync codes S₁, S₂, S₃, and S₄, respectively. Inthe ATSC standard published 16 Sep. 1995 the data segmentsynchronization code group consists of four symbols having successivevalues of +S, −S, −S and +S. In possible revisions of this standard thedata segment synchronization code group consists of six symbols thatexhibit asymmetry. One such code group consists of six symbols havingsuccessive values of +S, −S, −S, −S, +S and +S; another such code groupconsists of six symbols having successive values of +S, +S, −S, −S, −Sand +S; and other such code groups are the complements of the first twogroups. The invention can accommodate the use of any of these datasegment synchronization code groups.

FIG. 3A shows data segment L₁ as consisting of nothing after the datasegment sync code S₁, unless an optional replica S₃″ of a later-includeddata segment sync code S₃ is transmitted. In actuality, the data segmentL₁ is transmitted with an accompanying pilot signal which is ignored inFIG. 3A. Furthermore, in actuality post-ghosts of the data segment synccode S₁ and of the data previous thereto may extend 40 microseconds orso into the beginning of the 77.3-microsecond-long data segment L₁. Thefinal 37.3 microseconds or so of the data segment L₁ are presumablyabsent of post-ghosts of the data segment sync code S₁ and of the dataprevious thereto, which facilitates the detection of pre-ghosts of thedata segment sync code S₂ and data frame sync code in the ensuing datasegment L₂.

FIG. 3B shows a first ghost-cancellation reference signal 41 located inthe earlier portions of the data segment L₂, but after the data segmentsync code S₂. A second ghost-cancellation reference signal 42 begins1368 symbol epochs (two NTSC horizontal scan line durations) after thefirst GCR signal 41 begins and is complementary (opposite in sense ofamplitude modulation) to the first GCR signal 41. Since data segmentduration is 832 symbol epochs or so, the second GCR signal 42 beginslate in the data segment L₃, as shown in FIG. 3C, and finishes early inthe data segment L₄, as shown in FIG. 3D. FIG. 3D shows the data segmentL₄ finishing with the 12-symbol pre-code 43 used to resume the twelveindependent data codestreams prescribed by the ATSC standard published16 Sep. 1995.

Presuming data segment duration is 832 symbol epochs, on the 537^(th)symbol epoch of the data segment L₃ shown in FIG. 3C there begins arepetition S₂ of the data segment sync code S₂ at the beginning of thedata segment L₂ shown in FIG. 3B; and on the 537^(th) symbol epoch ofthe data segment L₄ shown in FIG. 3D there begins a repetition S₃′ ofthe data segment sync code S₃ at the beginning of the data segment L₃shown in FIG. 3C. The repetitions S₂′ and S₃′ of the data segment synccodes S₂ and S₃ cancel responses to the data segment sync codes S₂ andS₃ from the comb filter 30 in the VSB DTV receiver. Accordingly, thereis no data segment sync code S₂ in the comb filter 30 response tointrude upon the determination of pre-ghosts of the subtractivelycombined first and second GCR signals 41 and 42; and there is no datasegment sync code S₃ in the comb filter 30 response to intrude upon thedetermination of post-ghosts of the subtractively combined first andsecond GCR signals 41 and 42. Including the optional replica S₃″ of datasegment sync code S₃ in the 297^(th) through 300^(th) symbol epochs ofthe data segment L₁ cancels an earlier response to the data segment synccode S₃ in the comb filter 30 response so as not to intrude upon thedetermination of pre-ghosts of the subtractively combined first andsecond GCR signals 41 and 42 that are very advanced in time before theprincipal signal, so much so that inclusion of the replica S₃″ of datasegment sync code S₃ is probably not of much benefit.

The GCR signals 41 and 42 are a substantial number of symbols in length,to increase the sensitivity of the match filtering for detectinglow-energy ghosts. The length of these GCR signals is limited by thedesire that GCR signal 42 finish sufficiently before the 12-symbolpre-code 43 at the end of the data segment L₄ that post-ghostcalculations can reach for at least 40 microseconds (431 symbol epochs)without encountering background clutter. This limits the duration of GCRsignals 41 and 42 to around 685 symbol epochs. If the GCR signals 41 and42 were to be pseudo-random noise sequences, favored because of theirpronounced auto-correlation responses, PN511 sequences would be thelongest possible ones. Rather than using a PN511 sequence as the basisof the GCR signals 41 and 42, it is preferable to use two PN255sequences in cascade, which two PN255 sequences are orthogonal to eachother. A PN sequence that is orthogonal to another PN sequence hassubstantially zero cross-correlation with the other. Simply reversingthe order of a PN sequence is one way to generate another PN sequence,which two PN sequences are orthogonal to each other.

FIG. 3B shows the GCR signal 41 being composed of a PN255 sequence 411immediately followed by an orthogonal PN255* sequence 412. FIGS. 3C and3D show the GCR signal 42 being composed of a PN255 sequence 421immediately followed by an orthogonal PN255* sequence 422. The downwardarrows on the two sequences 411 and 412 and the upward arrows on the twosequences 421 and 422 are indicative of the complementary senses ofmodulation of the GCR signals 41 and 42. The selection of particularPN255 sequences for the GCR signals 41 and 42 is restricted by the needfor the data segment sync code S₄ to be subsumed in the orthogonalPN255* sequence 422.

Cascading two orthogonal PN255 sequences provides the option ofcalculating kernel weights based on just one of the PN255 sequenceswithout interference from the other. This option can be used to reducematch filter hardware costs in a less expensive, lower performance VSBDTV receiver. In a higher quality VSB DTV receiver this option can speedcalculations during rapid changes in selected channel.

FIGS. 4A, 4B, 4C and 4D illustrate that the portions of the datasegments L₁, L₂, L₃, and L₄ in the data frame header that are shown asfree of symbols in FIGS. 3A, 3B, 3C and 3D can in fact be occupied bysymbols without interfering with kernel weighting calculations by thecomputer 29, so long as the symbols are repeated so as to cancel fromthose portions of the comb filter 30 response that are used fordetermining the locations of pre-ghosts and post-ghosts. These symbolsmay be used to code program information used to screen the channels tobe viewed, for example. These symbols may provide for a frame start flagcode that is shorter in length than the training signal for the channelequalization filtering.

FIG. 4A shows a frame start flag code 44 composed of a pseudo-randomnoise pair-sequence 441 followed by a truncated repetition 442 thereofpositioned within the data segment L₁. A pseudo-random noisepair-sequence or “PNP sequence” is defined as a pseudo-random noisesequence where the symbol rate of the sequence is half normal symbolrate, so each symbol insofar as the PNP sequence is concerned is a pairof symbols insofar as the data transmission system or a regular PNsequence is concerned. A reason for PN pair sequences or “PNP sequences”being preferred by the inventors as frame-start flag codes is that matchfilter response to these sequences can be obtained even under conditionswhere symbol synchronization has not yet been perfected. This can speedup getting the VSB DTV signal receiver into operation. The PNP sequencescan provide the basis for sensitive phase adjustments of the symbolclock.

The PNP sequence 441 is a PNP127 sequence consisting of 127 pairs ofonce-repeated symbols of normal symbol rate, occupies 254 symbol epochs,and has successive values of +S, +S, −S, −S, +S and +S in the last sixof those symbol epochs. A duplicate 451 of the PNP sequence 441 starts1368 symbol epochs later than the PNP sequence 441 starts. The PNPsequence 451 starts within the data segment L₂, as shown in FIG. 4B, andfinishes with its last five symbols at the very beginning of datasegment L₃, as shown in FIG. 4C. This provides for the data segment synccode S₃ to be subsumed in the PNP sequence 451.

The PNP sequence 441 finishes 1368 symbol epochs earlier than the PNPsequence 451 starts, so the PNP sequence 441 finishes on the 301^(th)symbol epoch of data segment L₁ and consequently begins 254 symbolepochs earlier on the 48^(th) symbol epoch of data segment L₁. Thetruncated PNP sequence 441 starts on the 301^(st) symbol epoch of datasegment L₁ and finishes after the 431^(st) symbol epoch of data segmentL₁, but not later than the 522^(nd) symbol epoch of data segment L₁.

A second duplicate 461 of the PNP sequence 441 starts 1368 symbol epochslater than the PNP sequence 451 starts and so starts on the 288^(th)symbol epoch of data segment L₄, per the showing of FIG. 4D. The firstduplicate 451 of the PNP sequence 441 is immediately followed by a firstduplicate 452 of the truncated PNP sequence 442 in data segment L₃ asshown in FIG. 4C. The second duplicate 461 of the PNP sequence 441 isimmediately followed by a second duplicate 462 of the truncated PNPsequence 442 in data segment L₄ as shown in FIG. 4D.

The 12-symbol pre-code 43 at the finish of the data segment L₄ is shownin FIG. 4C as being preceded by a duplicate 12-symbol pre-code 431beginning in the data segment L₃ 1368 symbol epochs earlier than the12-symbol pre-code 43. The duplicate 12-symbol pre-code 431 cancels the12-symbol pre-code 43 in the comb filter 30 response to increase thetime that there is no background clutter to interfere with the computer29 calculation of post-ghosts. The duplicate 12-symbol pre-code 431 isshown in FIG. 4A as being preceded by another duplicate 12-symbolpre-code 432 beginning in the data segment L₁ 1368 symbol epochs earlierthan the 12-symbol pre-code 431. The duplicate 12-symbol pre-code 432cancels the 12-symbol pre-code 431 in the comb filter response tomaintain the time that there is no background clutter to interfere withthe computer 29 calculation of pre-ghosts.

A 24-symbol transmission mode code 47 is disposed in the data segment L₄shown in FIG. 4D; a duplicate 471 of the mode code 47 beginning 1368symbol epochs earlier than the mode code 47 begins is disposed in thedata segment L₃ shown in FIG. 4C; and a further duplicate 472 of themode code 47 beginning 2736 symbol epochs earlier than the mode code 47begins is disposed in the data segment L₁ shown in FIG. 4A. These24-symbol transmission mode codes 47, 471 and 472 can be delayed up tothe point where the 12-symbol pre-codes would be overlapped.Alternatively, 24-symbol transmission mode codes 47, 471 and 472 can beadvanced to precede the PNP sequences 441, 451 and 461, respectively, upto the point where the PN255* sequences 412 and 422 would be overlapped.In yet another variant the 24-symbol transmission mode code 472 can bedispensed with and the 24-symbol transmission mode codes 47 and 471advanced to positions prior to the data segment sync codes S₂′ and S₂,respectively, up to the point where the sequences 412 and 422 would beoverlapped.

A problem with the data frame header shown in FIGS. 3A, 3B, 3C and 3D isthat the repetitions S₂′ and S₃′ of the data segment sync codes S₂ andS₃ violate the mandate of the ATSC standard published 16 Sep. 1995 thata code group consisting of four symbols having successive values of +S,−S, −S and +S not be repeated at a 832-symbol interval unless it ispositioned at the beginning of the data segment to be used as datasegment synchronization code. This problem is also found in the dataframe header shown in FIGS. 4A, 4B, 4C and 4D. This problem can beavoided simply, by omitting the data segment sync code S₂ at thebeginning of the data segment L₂ and by omitting the repetition S₂′ ofthat data segment sync code within the succeeding data segment L₃.Alternatively, this problem can be avoided simply, by omitting the datasegment sync code S₃ at the beginning of the data segment L₃, byomitting the repetition S₃′ of the data segment sync code S₃ within thesucceeding data segment L₄, and by omitting any duplicate S₃′ of datasegment sync code S₃ within the data segment L₁. The circuitry used fordetecting data segment sync code normally includes provision foraccommodating the non-occurrence of such code at the beginning of onedata segment.

FIGS. 5A, 5B, 5C and 5D show a variant of the data frame header shown inFIGS. 3A, 3B, 3C and 3D that avoids the violation of the ATSC standardpublished 16 Sep. 1995 in a more elegant manner by subsuming the datasegment sync code S₃ within the first four symbol epochs of a PN255sequence 411′, which replaces the PN255 sequence 411 and is generated bycylindrical rotation of the PN255 sequence 411 until a data segment synccode heads the PN sequence. The PN255 sequence 421 is replaced by aPN255 sequence 421′, which is complementary to the PN255 sequence 411′.Since the PN255 sequence 421′ is complementary to the PN255 sequence411′, the data segment sync code S₃ is not repeated in the beginning ofthe PN255 sequence 421′.

Along the same lines, FIGS. 6A, 6B, 6C and 6D show a variant of the dataframe header shown in FIGS. 4A, 4B, 4C and 4D that avoids the violationof the ATSC standard published 16 Sep. 1995, by subsuming the datasegment sync code S₃ within the PN255 sequence 411′. The data frameheaders shown in FIGS. 5A, 5B, 5C and 5D and in FIGS. 6A, 6B, 6C and 6Dpermit ready correction of pre-ghosts up to 37.3 microseconds in advanceof the principal signal without interference of post-ghosts delayed asmuch as 40 microseconds from the principal signal, then subsequentcorrection of post-ghosts delayed 20.9 to 58.2 microseconds from theprincipal signal without interference of pre-ghosts or other backgroundclutter, then subsequent correction of pre-ghosts up to 56.4microseconds in advance of the principal signal without interference ofpost-ghosts delayed more than 20.9 microseconds from the principalsignal, then subsequent correction of post-ghosts delayed 1.8 to 20.9microseconds from the principal signal without interference ofpre-ghosts, then subsequent correction of pre-ghosts up to 75.5microseconds in advance of the principal signal without interference ofpost-ghosts delayed more than 1.8 microseconds from the principalsignal, then subsequent correction of post-ghosts delayed less than 1.8microseconds from the principal signal, and then correction ofmicro-ghosts in a final equalization step. The later ghost-cancellationand equalization steps may be performed more expeditiously usingdiscrete Fourier transform methods.

FIGS. 7A, 7B, 7C and 7D show a variant of the data frame header shown inFIGS. 5A, 5B, 5C and 5D that increases the capability for suppressingpost-ghosts delayed over post-ghosts delayed over 58.2 microseconds fromthe principal signal without interference from background clutter, whilereducing the capability for suppressing pre-ghosts far in advance ofprincipal signal. The first GCR signal 41 and second GCR signal 42 areadvanced in time so that the first GCR signal 41 commences in the firstdata segment L₁ more than 58.2 microseconds into the data segment. ThePN255 sequence 413 is replaced by a PN255 sequence 414 generated bycylindric rotation of the PN255 sequence 413 until the data segment synccode is positioned within the PN sequence so as to fall at the outset ofthe second data segment L₂. The PN255 sequence 423 is replaced by aPN255 sequence 424 that is complementary to the PN255 sequence 414.

Along the same lines, FIGS. 8A, 8B, 8C and 8D show a variant of the dataframe header shown in FIGS. 6A, 6B, 6C and 6D that increases thecapability for suppressing post-ghosts delayed over post-ghosts delayedover 58.2 microseconds from the principal signal without interferencefrom background clutter, while reducing the capability for suppressingpre-ghosts far in advance of principal signal. Advancing the GCR signals41 and 42 so that the first GCR signal 41 commences in the first datasegment L₁ lifts the restriction of the duration of each of these GCRsignals from around 685 symbol epochs towards 1023 symbol epochs, sothat the use of PN sequences of longer length and better noise-rejectioncapability can be contemplated.

FIGS. 9A, 9B, 9C and 9D show a data frame header in which a first GCRsignal 51 comprises an initial PN511 sequence 511, which commences inthe first data segment L₁ and has its last four symbols at the beginningof the second data segment L₂ so as to provide the data segment synccode S₂, and a final PN511 sequence 512, which is disposed entirelywithin the second data segment L₂. A second GCR signal 52 begins 1368symbol epochs after the first GCR signal 51 begins. The second GCRsignal 52 comprises an initial PN511 sequence 521, which occupies the29^(th) through 540^(th) symbol epochs of the third data segment L₃, anda final PN511 sequence 522, which begins in the third data segment L₂and finishes in the fourth data segment L₄. The data segment sync codeS₄, at the outset of the fourth data segment L₄, is subsumed in thefinal PN511 sequence 522. A repetition 531 of the beginning of the PN511sequence 521 starts 1368 symbol epochs after the PN511 sequence 521starts.

FIG. 9B shows a first repetition S₁′ of the data segment sync code S₁that begins 1368 symbol epochs after the data segment sync code S₁begins, and FIG. 9D shows a second repetition S₁″ of the data segmentsync code S₁ that begins 1368 symbol epochs after the first repetitionS₁′ of the data segment sync code S₁ begins. FIG. 9 D also shows arepetition S₃′ of the data segment sync code S₃ that begins 1368 symbolepochs after the data segment sync code S₃ begins, and FIG. 9A showsanother duplicate S₃″ of the data segment sync code S₃ that begins 1368symbol epochs before the data segment sync code S₃ begins. Accordingly,the comb filter 30 used for suppressing artifacts of co-channel NTSCinterference in the baseband symbol code received by VSB DTV signalreceiver is not responsive to the data segment sync code S₁ nor to thedata segment sync code S₃.

FIG. 9D shows the 24-symbol mode code 47 located just prior to therepetition S₃′ of the data segment sync code S₃. This location of the24-symbol mode code 47 or an alternative location just after the secondrepetition S₁″ of the data segment sync code S₁ leaves room forinsertion of a 272-symbol frame-start flag code next to the mode codeand between the code segments S₁″ and S₃′, if a designer desires. APNP127 sequence will fit into this space with a little bit of room forrepetition.

FIG. 10 diagrams a data frame of 1252 data segments, with timeprogressing left to right and top to bottom. The initial four datasegments in the data frame are a data header 61 composed of datasegments L₁, L₂, L₃, and L₄ as described above. The last 1248 datasegments begin with respective four-symbol data synchronization codes62, as do at least certain ones of the data segments L₁, L₂, L₃, and L₄in the data header 61. The respective four-symbol data synchronizationcodes 62 in the last 1248 data segments are followed by error-correctioncoded data payload 63. Payload data rate using the 1252-data-segmentdata frame with four-data-segment header is 19.28 Mbps, the same as withthe current ATSC standard. The 1252-data-segment data frame with4-data-segment data field synchronization code shown in FIG. 10 replacesfour of the 313-data-segment data fields with respective data fieldsynchronization codes per the ATSC DTV standard published 16 Sep. 1995.The one-tenth of a second interval between data frame synchronizationcode headers supplies training signal for correcting the DTV receiverequalization frequently enough to permit successive tuning of variouschannels without hesitation in acquiring each new channel beingnoticeable to a person viewing the televised images.

A problem with the 1252-data-segment frame of FIG. 10 is that 5.17 dataframes are co-extensive with a 15-image-frame group of pictures (GOP)that is of 0.5005005005005 second duration. This presents problems incueing and editing of digital television recording tapes on a digitalframe basis. It is preferable that insofar as possible the15-image-frame GOP be co-extensive in time with five data frames. Thisalso provides for a 3-, 6-, 9- or 12-image-frame GOP being coextensivein time with an integral number of data frames. A data frame with 1294or 1295 of the 832-symbol data segments therein is much closer to whatis desired. At times, initialization of the adaptive equalizer can takeno shorter a time than the duration of 1079 data segments (about1/12^(th) second) required worst case for acquiring the training signaland will take somewhat more time for calculation of the kernel weightsfor the adaptive equalizer. This is adequate speed for initialization ofthe adaptive equalizer response.

FIG. 11 diagrams a data frame of 1294 data segments, with timeprogressing left to right and top to bottom. The initial four datasegments in the data frame are a data header 61 composed of datasegments L₁, L₂, L₃, and L₄ as described above. The last 1290 datasegments begin with respective four-symbol data synchronization codes62, as do at least certain ones of the data segments L₁, L₂, L₃, and L₄in the data header 611. The respective four-symbol data synchronizationcodes 62 in the last 1290 data segments are followed byerror-correction-coded data payload 64.

The use of a data frame with 1294 data segments therein makes itdesirable to redesign the convolutional interleaving in the transmittedVSB DTV signal and the convolutional de-interleaver 16 in the VSB DTVreceiver of FIG. 1. Re-design is desirable because the 52-data-segmentinterleaving depth that fits exactly six times into the 312-data-segmentdata field (sans header) in the 1995 ATSC standard does not fit anintegral number of times into the 1290-data-segmenterror-correction-coded data payload 64. It is desirable that theinterleaving depth exactly fit an integral number of times into the dataframe, to facilitate switching between signal sources at the DTVtransmitter. A 43-data-segment interleaving depth fits exactly 30 timesinto the 1290-data-segment data frame (sans header). There can be lessstorage in the de-interleaver 16 than required in a de-interleaver forVSB DTV signal conforming to the 1995 ATSC standard. Impulse noise mustbe less than 43 symbol epochs duration to be treated as single errors bythe R-S decoder 20, but impulse noise is usually shorter than this fourmicrosecond interval if the RF and IF amplifiers in the DTV signalreceiver are designed reasonably well. Alternatively, an 86-data-segmentinterleaving depth fits exactly 15 times into the 1290-data-segmenterror-correction-coded data payload 64. While more buffer memory isrequired for the greater interleaving depth, error correction forlong-duration impulse noise is improved. In yet another alternative, a215-data-segment interleaving depth fits exactly six times into the1290-data-segment error-correction-coded data payload 64, but still morebuffer memory is required for the even greater interleaving depth.

Payload data rate using the 1252-data-segment data frame withfour-data-segment header is 19.28 Mbps, the same as with the currentATSC standard. Using a 1294-data-segment data frame with afour-data-segment header increases payload data rate slightly above the19.28 Mbps achieved using the 1252-data-segment data frame withfour-data-segment header. A variant using a 1295-data-segment data framewith a five-data-segment header reduces payload data rate to 19.267Mbps, but there is closer temporal tracking of image frames and dataframes.

The data segment synchronization codes beginning data segments in thecurrent ATSC standard are overhead that is unnecessary for awell-designed data communications receiver. The data segmentsynchronization codes were used in primitive DTV receiver designs forperiodically correcting sampling clock rates to concur with symbol rate.Sampling clock rate adjustments are preferably made on a continuousbasis using spectral line filtering of intermediate-frequency signalenvelope to recover a symbol rate subharmonic and synchronizing samplingclock rate to the recovered symbol rate subharmonic. The interpositionof the data segment synchronization codes into trellis codingundesirably complicates the trellis decoding procedures in the receiver.

FIG. 12 diagrams a data frame of 1302 data segments, with timeprogressing left to right and top to bottom. Data segmentsynchronization codes are discarded to leave 828-symbol data segments ina data header 65 and in an error-correction-coded data payload 66. Thegeneral nature of the data header 65 is similar to that of the dataheader 61. Payload data rate using these 1302-data-segment data frameswith four-data-segment headers is 19.38 Mbps. The convolutionalinterleaving in the transmitted VSB DTV signal and the convolutionalde-interleaver 16 in the FIG. 1 VSB DTV signal receiver are re-designed.A 59-data-segment interleaving depth fits exactly 22 times into the 1298data segments of the data payload 66. Alternatively, a 118-data-segmentinterleaving depth fits exactly eleven times into the 1298 datasegments.

In a variant of the scheme illustrated in FIG. 12 a data frame of 1301data segments with a five-data-segment header is used, leaving 1296 datasegments per frame for trellis-coded data. Payload data rate using these1301-data-segment data frames with five-data-segment headers is 19.36Mbps. A 54-data-segment interleaving depth fits exactly 24 times intothe 1296 data segments of the 1301-data-segment error-correction-codeddata payload. Alternatively, a 72-data-segment interleaving depth fitsexactly 18 times into the 1296 data segments. In still furtheralternatives, a 108-data-segment interleaving depth fits exactly 12times into the 1296 data segments, or a 144-data-segment interleavingdepth fits exactly 9 times into the 1296 data segments. The 54-, 72- and108-data-segment interleaving depths lend themselves particularly totime-division multiplexing of six reduced-resolution television signalsin each data frame.

While ghost-reference signals composed of just pairs of orthogonal PNsequences have been described, ghost-reference signals with a largernumber of orthogonal PN sequences exist, which can be used in furtherembodiments of the invention. Such ghost-reference signals can, forexample, each be composed of cascaded complementary sequences asdescribed in U.S. Pat. Nos. 5,341,177 and 5,361,102 issued to Sumit Roy,C. B. Patel and J. Yang on 23 Aug. 1994 and on 1 Nov. 1994,respectively, and respectively entitled “SYSTEM TO CANCEL GHOSTSGENERATED BY MULTIPATH TRANSMISSION OF TELEVISION SIGNAL” and “SYSTEM TOCANCEL GHOSTS IN NTSC TELEVISION TRANSMISSION”. The advantage of aghost-reference signal composed of cascaded complementary sequences ascompared to a ghost-reference signal consisting of a PN sequence is thatit contains all spectral frequency components with substantially equalenergies.

Trellis coding in a VSB DTV signal per the 1995 ATSC standard extendscontinuously from frame to frame. This complicates switching betweensignal sources at the broadcast transmitter or in the studio, especiallywhen the DTV signal time-division-multiplexes a plurality of programsand services. A better practice is to restart trellis coding at thebeginning of each block of data segments that are convolutionallyinterleaved with each other, or at least at the beginning of the firstblock of such data segments in each data frame. The trellis coding ispreferably restarted after the data frame header using a prescribedprecode, so precode information does not have to be included in thetransmitted signal.

1. A data signal receiver for an electromagnetic wave signal including apilot carrier and vestigial sideband modulation of a suppressed carrierof the same frequency and phase as said pilot carrier, said vestigialsideband modulation being in accordance with a baseband signal having auniform symbol rate substantially 684 times the horizontal scan linerate of an NTSC television signal that is apt to accompany saidelectromagnetic wave signal as a co-channel interfering signal, saidbaseband signal composed of consecutive data segments each consisting ofa prescribed integral number of symbol epochs, said consecutive datasegments being divided into contiguous data frames each consisting of aprescribed integral number M of contiguous ones of said data segments,each said data frame characterized by beginning with a data frame headerincluding a plurality N in number of contiguous ones of said datasegments and concluding with a plurality (M-N) in number of said datasegments including consecutive multi-level symbols used for transmittingdata, said data segments used for special purposes in each said dataframe including a first ghost-cancellation reference signal and a secondghost-cancellation reference signal at a prescribed time intervalthereafter, which said first and second ghost-cancellation referencesignal exhibit respective variations that are complementary to eachother, said data signal receiver comprising: circuitry for selectingsaid electromagnetic wave signal, converting the frequencies of saidelectromagnetic wave signal after its selection, and amplifying saidelectromagnetic wave signal after its selection and conversion infrequency; circuitry for synchrodyning said electromagnetic wave signalto baseband after its selection, conversion in frequency andamplification and supplying digitized samples of a baseband signalresulting from synchrodyning said electromagnetic wave signal tobaseband; an adaptive equalizer for receiving said samples of a basebandsignal resulting from synchrodyning said electromagnetic wave signal tobaseband, and supplying an equalizer response to those received samplesas weighted by kernel weights that are electrically adjustable;circuitry for regenerating transmitted data from said equalizerresponse; a comb filter for differentially delaying said equalizerresponse, so said first ghost-cancellation reference signal in the moredelayed equalizer response occurs simultaneously with said secondghost-cancellation reference signal in the less delayed equalizerresponse, and subtractively combining said more delayed equalizerresponse and said less delayed equalizer response to generate a combfilter response; and a computer responsive to selected portions of saidcomb filter response including the result of subtractively combiningsaid first and second ghost-cancellation reference signals, forperforming initial electrical adjustments of the kernel weights of saidadaptive equalizer whenever said data signal receiver is initiallyoperated after a time of in operation or whenever said electromagneticwave signal is initially selected.
 2. The data signal receiver of claim1, wherein said comb filter differentially delays said equalizerresponse by substantially 1368 symbol epochs to generate said moredelayed equalizer response and said less delayed equalizer response forbeing subtractively combined to generate said comb filter response. 3.The data signal receiver of claim 1, in which during continued operationof said said data signal receiver said computer continues toelectrically adjust the kernel weights of said adaptive equalizerresponsive to selected portions of said comb filter response includingthe result of subtractively combining said first and secondghost-cancellation reference signals.
 4. The data signal receiver ofclaim 3, wherein said comb filter differentially delays said equalizerresponse by substantially 1368 symbol epochs to generate said moredelayed equalizer response and said less delayed equalizer response forbeing subtractively combined to generate said comb filter response. 5.The data signal receiver of claim 1, in which during continued operationof said said data signal receiver said computer electrically adjusts thekernel weights of said adaptive equalizer responsive to said comb filterresponse on a decision-directed basis.
 6. The data signal receiver ofclaim 5, wherein said comb filter differentially delays said equalizerresponse by substantially 1368 symbol epochs to generate said moredelayed equalizer response and said less delayed equalizer response forbeing subtractively combined to generate said comb filter response. 7.An electromagnetic wave signal received and processed by a televisionsignal receiver, said signal comprising vestigial sideband modulation ofa suppressed carrier in accordance with a baseband signal having auniform baud rate or symbol rate substantially 684 times the horizontalscan line rate of an NTSC television signal that is apt to accompanysaid electromagnetic wave signal as a co-channel interfering signal,said baseband signal composed of consecutive data segments eachconsisting of a prescribed integral number of symbol epochs, saidconsecutive data segments being divided into contiguous data frames eachconsisting of a prescribed integral number M of contiguous ones of saiddata segments, each said data frame characterized by beginning with adata frame header including a plurality N in number of contiguous onesof said data segments and concluding with a plurality (M-N) in number ofsaid data segments including consecutive multi-level symbols used fortransmitting data, said data frame header in each said data frameincluding a first ghost-cancellation reference signal and a secondghost-cancellation reference signal beginning substantially 1368 symbolepochs later than said first ghost-cancellation reference signal, whichsaid first and second ghost-cancellation reference signal exhibitrespective variations that are complementary to each other.
 8. Theelectromagnetic wave signal of claim 7, wherein each of said plurality(M-N) in number of said data segments composed of consecutivemulti-level symbols used for transmitting data begins with a four-symboldata-segment-synchronizing code.
 9. The electromagnetic wave signal ofclaim 8, wherein said number M has a value equal to 1254 and said numberN equals four.
 10. The electromagnetic wave signal of claim 8, whereinsaid number M has a value equal to 1294 and said number N equals four.11. The electromagnetic wave signal of claim 8, wherein said number Mhas a value equal to 1295 and said number N equals five.
 12. Theelectromagnetic wave signal of claim 7, wherein said number M has avalue equal to 1302 and said number N equals four.
 13. Theelectromagnetic wave signal of claim 7, wherein said number M has avalue equal to 1301 and said number N equals five.
 14. Theelectromagnetic wave signal of claim 7, wherein said firstghost-cancellation reference signal is composed of a plurality of PNsequences that are orthogonal to each other and contain equal numbers ofsymbols.
 15. The electromagnetic wave signal of claim 14, wherein saidnumber N has a value at least four, and wherein said firstghost-cancellation reference signal begins before the end of the firstdata segment of each data frame.
 16. The electromagnetic wave signal ofclaim 14, wherein said number N has a value at least four, and whereinsaid first ghost-cancellation reference signal begins at the beginningof the second data segment of each data frame.
 17. The electromagneticwave signal of claim 14, wherein said number N has a value at leastfour, and wherein said first ghost-cancellation reference signal beginsa few symbol epochs after the beginning of the second data segment ofeach data frame.
 18. The electromagnetic wave signal of claim 7, whereina frame start signal is included in the first data segment of each dataframe, said first ghost-cancellation reference signal begins after saidframe start signal, and said second ghost-cancellation reference signalbegins substantially 1368 symbol epochs after said firstghost-cancellation reference signal begins.
 19. The electromagnetic wavesignal of claim 18, wherein said number N has a value at least four, andwherein said first ghost-cancellation reference signal begins before theend of the first data segment of each data frame.
 20. Theelectromagnetic wave signal of claim 18, wherein said number N has avalue at least four, and wherein said first ghost-cancellation referencesignal begins at the beginning of the second data segment of each dataframe.
 21. The electromagnetic wave signal of claim 18, wherein saidnumber N has a value at least four, and wherein said firstghost-cancellation reference signal begins a few symbol epochs after thebeginning of the second data segment of each data frame.
 22. Theelectromagnetic wave signal of claim 18, wherein said frame start signalcomprises a pseudo-random noise sequence with a baud rate or symbol ratesubstantially 342 times the horizontal scan line rate of an NTSCtelevision signal.
 23. The electromagnetic wave signal of claim 22,wherein said pseudo-random noise sequence with a baud rate or symbolrate substantially 342 times the horizontal scan line rate of an NTSCtelevision signal is succeeded within said frame start signal by asignal corresponding to at least the initial portion of thatpseudo-random noise sequence.
 24. An electromagnetic wave signalreceived and processed by a television signal receiver, said signalcomprising vestigial sideband modulation of a suppressed carrier inaccordance with a baseband signal having a uniform baud rate or symbolrate, said baseband signal composed of consecutive data segments eachconsisting of a prescribed integral number of symbol epochs, saidconsecutive data segments being divided into contiguous data frames eachconsisting of a prescribed integral number M of contiguous ones of saiddata segments, each said data frame characterized by beginning with aplurality N in number of said data segments used as a data frame headerand concluding with a plurality (M-N) in number of said data segmentsthat include consecutive multi-level symbols used for transmitting data,said data frame header in each said data frame including a respectiveghost-cancellation reference signal that is composed of a plurality ofPN sequences that are orthogonal to each other.
 25. The electromagneticwave signal of claim 24, wherein in each said data frame header saidghost-cancellation reference signal exhibits variation that iscomplementary to variation exhibited by a preceding otherghost-cancellation reference signal in the same said data frame header.26. An electromagnetic wave signal received and processed by atelevision signal receiver, said signal comprising vestigial sidebandmodulation of a suppressed carrier in accordance with a baseband signalhaving a uniform symbol rate, said baseband signal composed ofconsecutive data segments each consisting of a prescribed integralnumber of symbol epochs, said consecutive data segments being dividedinto contiguous data frames each consisting of a prescribed integralnumber M of contiguous ones of said data segments, each said data framecharacterized by beginning with a data frame header including aplurality N in number of contiguous ones of said data segments andconcluding with a plurality (M-N) in number of said data segmentsincluding consecutive multi-level symbols used for transmitting data,said data segments each beginning with a respective data segmentsynchronization code of a similar prescribed character, said data frameheader in each said data frame including a respective ghost-cancellationreference signal that begins in one data segment of said data frameheader and finishes in the next-occurring data segment of said dataframe header, said respective data segment synchronization code for saidnext data segment of said data frame header being subsumed in saidrespective ghost-cancellation reference signal that finishes therein.27. The electromagnetic wave signal of claim 26, wherein in each saiddata frame header said ghost-cancellation reference signal exhibitsvariation that is complementary to variation exhibited by anotherghost-cancellation reference signal in the same said data frame header.28. The electromagnetic wave signal of claim 27, wherein in each saiddata frame header said ghost-cancellation reference signal begins in athird-occurring data segment of said data frame header, finishes in afourth-occurring data segment of said data frame header, and exhibitsvariation that is complementary to variation exhibited by anotherghost-cancellation reference signal beginning after said respective datasegment synchronization code in a second-occurring data segment of saiddata frame header.
 29. The electromagnetic wave signal of claim 28,wherein in each said data frame header said ghost-cancellation referencesignal begins in a third-occurring data segment of said data frameheader, finishes in a fourth-occurring data segment of said data frameheader, and exhibits variation that is complementary to variationexhibited by another ghost-cancellation reference signal beginning atthe outset of a second-occurring data segment of said data frame header,said respective data segment synchronization code for said second datasegment of said data frame header being subsumed in said otherghost-cancellation reference signal therewithin.
 30. The electromagneticwave signal of claim 27, wherein in each said data frame header saidghost-cancellation reference signal begins in a third-occurring datasegment of said data frame header, finishes in a fourth-occurring datasegment of said data frame header, and exhibits variation that iscomplementary to variation exhibited by another ghost-cancellationreference signal beginning in a first-occurring data segment of saiddata frame header and finishing in a second-occurring data segment ofsaid data frame header, said respective data segment synchronizationcode for said second data segment of said data frame header beingsubsumed in said other ghost-cancellation reference signal.
 31. Abaseband digital signal received and processed by a television signalreceiver, said signal having a uniform symbol rate substantially 684times the horizontal scan line rate of an NTSC television signal that isapt to accompany said electromagnetic wave signal as a co-channelinterfering signal, wherein said baseband signal is composed ofconsecutive data segments each consisting of a prescribed integralnumber of symbol epochs, said consecutive data segments being dividedinto contiguous data frames each consisting of a prescribed integralnumber M of contiguous ones of said data segments, each said data framecharacterized by beginning with a plurality N in number of said datasegments used as a data frame header and concluding with a plurality(M-N) in number of said data segments composed of consecutivemulti-level symbols used for transmitting data, said data frame headerin each said data frame including a first ghost-cancellation referencesignal and a second ghost-cancellation reference signal beginningsubstantially 1368 symbol epochs later than said firstghost-cancellation reference signal, which said first and secondghost-cancellation reference signal exhibit respective variations thatare complementary to each other.
 32. The baseband digital signal ofclaim 31, wherein each of said plurality (M-N) in number of said datasegments composed of consecutive multi-level symbols used fortransmitting data begins with a four-symbol data-segment-synchronizingcode.
 33. The baseband digital signal of claim 32, wherein said number Mequals 1252 and said number N equals four.
 34. The baseband digitalsignal of claim 32, wherein said number M equals 1294 and said number Nequals four.
 35. The baseband digital signal of claim 32, wherein saidnumber M equals 1295 and said number N equals five.
 36. The basebanddigital signal of claim 31, wherein said number M equals 1302 and saidnumber N equals four.
 37. The baseband digital signal of claim 31,wherein said number M equals 1301 and said number N equals five.
 38. Thebaseband digital signal of claim 31, wherein said firstghost-cancellation reference signal is composed of a plurality of PNsequences that are orthogonal to each other and contain equal numbers ofsymbols.
 39. The baseband digital signal of claim 38, wherein saidnumber N is at least four, and wherein said first ghost-cancellationreference signal begins before the end of the first data segment of eachdata frame.
 40. The baseband digital signal of 38, wherein said number Nis at least four, and wherein said first ghost-cancellation referencesignal begins at the beginning of the second data segment of each dataframe.
 41. The baseband digital signal of claim 38, wherein said numberN is at least four, and wherein said first ghost-cancellation referencesignal begins a few symbol epochs after the beginning of the second datasegment of each data frame.
 42. The baseband digital signal of claim 31,wherein a frame start signal is included in the first data segment ofeach data frame, said first ghost-cancellation reference signal beginsafter said frame start signal, and said second ghost-cancellationreference signal begins substantially 1368 symbol epochs after saidfirst ghost-cancellation reference signal begins.
 43. The basebanddigital signal of claim 42, wherein said number N is at least four, andwherein said first ghost-cancellation reference signal begins before theend of the first data segment of each data frame.
 44. The basebanddigital signal of claim 42, wherein said number N is at least four, andwherein said first ghost-cancellation reference signal begins at thebeginning of the second data segment of each data frame.
 45. Thebaseband digital signal of claim 42, wherein said number N is at leastfour, and wherein said first ghost-cancellation reference signal beginsa few symbol epochs after the beginning of the second data segment ofeach data frame.
 46. The baseband digital signal of claim 42, whereinsaid frame start signal comprises a pseudo-random noise sequence with asymbol rate substantially 342 times the horizontal scan line rate of anNTSC television signal.
 47. The baseband digital signal of claim 46,wherein said pseudo-random noise sequence with a symbol ratesubstantially 342 times the horizontal scan line rate of an NTSCtelevision signal is succeeded within said frame start signal by asignal corresponding to at least the initial portion of thatpseudo-random noise sequence.
 48. A baseband signal received andprocessed by a television signal receiver, said baseband signal having auniform symbol rate and being composed of consecutive data segments eachconsisting of a prescribed integral number of symbol epochs, saidconsecutive data segments being divided into contiguous data frames eachconsisting of a prescribed integral number M of contiguous ones of saiddata segments, each said data frame characterized by beginning with adata frame header including a plurality N in number of contiguous onesof said data segments and concluding with a plurality (M-N) in number ofsaid data segments including consecutive multi-level symbols used fortransmitting data, said data frame header in each said data frameincluding a respective ghost-cancellation reference signal that iscomposed of a plurality of PN sequences that are orthogonal to eachother.
 49. The baseband signal of claim 48, wherein in each said dataframe header said ghost-cancellation reference signal exhibits variationthat is complementary to variation exhibited by a preceding otherghost-cancellation reference signal in the same said data frame header.50. A baseband signal received and processed by a television signalreceiver, said baseband signal having a uniform symbol rate, saidbaseband signal and being composed of consecutive data segments eachconsisting of a prescribed integral number of symbol epochs, saidconsecutive data segments being divided into contiguous data frames eachconsisting of a prescribed integral number M of contiguous ones of saiddata segments, each said data frame characterized by beginning with adata frame header including a plurality N in number of contiguous onesof said data segments and concluding with a plurality (M-N) in number ofsaid data segments including consecutive multi-level symbols used fortransmitting data, said data segments each beginning with a respectivedata segment synchronization code of a similar prescribed character,said data frame header in each said data frame including a respectiveghost-cancellation reference signal that begins in one data segment ofsaid data frame header and finishes in the next-occurring data segmentof said data frame header, said respective data segment synchronizationcode for said next data segment of said data frame header being subsumedin said respective ghost-cancellation reference signal that finishestherein.
 51. The baseband signal of claim 50, wherein in each said dataframe header said ghost-cancellation reference signal exhibits variationthat is complementary to variation exhibited by anotherghost-cancellation reference signal in the same said data frame header.52. The baseband signal of claim 51, wherein in each said data frameheader said ghost-cancellation reference signal begins in athird-occurring data segment of said data frame header, finishes in afourth-occurring data segment of said data frame header, and exhibitsvariation that is complementary to variation exhibited by anotherghost-cancellation reference signal beginning after said respective datasegment synchronization code in a second-occurring data segment of saiddata frame header.
 53. The baseband signal of claim 52, wherein in eachsaid data frame header said ghost-cancellation reference signal beginsin a third-occurring data segment of said data frame header, finishes ina fourth-occurring data segment of said data frame header, and exhibitsvariation that is complementary to variation exhibited by anotherghost-cancellation reference signal beginning at the outset of asecond-occurring data segment of said data frame header, said respectivedata segment synchronization code for said second data segment of saiddata frame header being subsumed in said other ghost-cancellationreference signal therewithin.
 54. The baseband signal of claim 51,wherein in each said data frame header said ghost-cancellation referencesignal begins in a third-occurring data segment of said data frameheader, finishes in a fourth-occurring data segment of said data frameheader, and exhibits variation that is complementary to variationexhibited by another ghost-cancellation reference signal beginning in afirst-occurring data segment of said data frame header and finishing ina second-occurring data segment of said data frame header, saidrespective data segment synchronization code for said second datasegment of said data frame header being subsumed in said otherghost-cancellation reference signal.